Integrated electronic device comprising a temperature sensor and sensing method

ABSTRACT

A method of sensing a temperature includes providing a voltage to reverse bias a PN junction of a junction diode. The PN junction has a junction capacitance. The method includes providing a reverse bias voltage change across the PN junction and detecting a value of the junction capacitance in response to the reverse bias voltage change. The value of the junction capacitance is a function of a temperature of the PN junction. An output signal is generated based on the detected junction capacitance, where the output signal indicates a temperature of an environment containing the junction diode.

BACKGROUND Technical Field

The present disclosure relates to an integrated electronic devicecomprising a temperature sensor and to the relevant sensing method.

Description of the Related Art

As is known, temperature sensors have a plurality of applications. Forinstance, they may be stand-alone components, which supply at output thetemperature value of an environment. In addition, they may be acomponent of a more complex system, which includes other elements, theperformance whereof varies with temperature. These variations arefrequently undesirable so that it is useful to detect the existingtemperature and compensate for the performance variations and make themindependent of temperature. Also when the performance variation withtemperature is the desired effect of the complex system, frequently itis in any case useful to have direct information on the local absolutetemperature value.

Temperature sensors are built in very different ways, in particularaccording to the application and to whether they are of a stand-alonetype or they are integrated in a more complex system. In the formercase, in fact, frequently no problems of dimensions exist, and simplerbut more cumbersome solutions may be used, whereas in the latter casethe possibility of an integrated implementation with the othercomponents of the system, in addition to the dimensions and consumption,may be important.

In case of temperature sensors integrated in an electronic circuit, itis known to exploit the variability with temperature of the base-emittervoltage of bipolar transistors. In fact, it is well known that thisvoltage has a variation of some millivolts per degree centigrade. Bydetecting the variation of voltage with a sensing circuit and amplifyingit, with appropriate algorithms it is possible to determine the localtemperature within the electronic circuit. This solution, albeitextremely widely adopted, is not free from disadvantages, due forexample to the need of implementing bipolar components for MOStechnology circuits and/or to the high consumption of the temperaturesensor and the associated components, for example of conditioning andamplification stages associated to the temperature sensor. Furthermore,this solution has the disadvantage of giving rise to high noise, whichmay be disadvantageous in some applications. On the other hand, theknown solutions have consumption levels that are the higher, the lowerthe level of maximum accepted noise. Not least, this solution does notalways solve the problem since the base-emitter voltage read isgenerally compared with a reference value, generated through a differentstage, such as a band-gap circuit, which in turn may vary withtemperature. This behavior introduces an error in the output signal, sothat the temperature value read may not have the desired precision.

In some known solutions, the sensing circuit comprises a resistivebridge for compensating for the temperature dependence in the referenceelement or circuit. However, also this solution is not free fromdisadvantages in so far as it introduces an undesirable consumptionlevel.

More innovative solutions comprise, for example, the use of MEMS(Micro-Electrical-Mechanical System) technologies that enable creationof elements that may undergo mechanical deformation as the temperaturevaries (see, for example, “A Micromachined Silicon CapacitiveTemperature Sensor for Radiosonde Applications” by Hong-Yu Ma, Qing-AnHuang, Ming Qin, Tingting Lu, E-ISBN: 978-1-4244-5335-1/09, 2009 IEEE).Other known solutions are based upon the use of new materials (see, forexample, “High-performance bulk silicon interdigital capacitivetemperature sensor based on graphene oxide” by Chun-Hua Cai and MingQin, ELECTRONICS LETTERS, 28 Mar. 2013 Vol. 49 No. 7, ISSN: 0013-5194).

These solutions are, however, difficult to integrate in digital systemsand thus not universally applicable.

BRIEF SUMMARY

One aim of the present disclosure is to provide a temperature sensorthat overcomes the drawbacks of the prior art.

According to the present disclosure, an integrated electronic deviceexploits the fact that a reverse biased PN junction has an equivalentcapacitance variable in a known way with temperature. This capacitancemay be compared with a reference capacitance provided to have anegligible dependence upon temperature. A known sensing circuit, forexample a switched capacitor operational amplifier, may then detect thecapacitance variation with temperature and output a voltage that variesdirectly with the capacitance variation.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

For a better understanding of the present disclosure preferredembodiments thereof are now described, purely by way of non-limitingexample, with reference to the attached drawings, wherein:

FIG. 1 shows the plot of the junction capacitance C_(j) of a junctiondiode as a function of the voltage applied V_(d);

FIG. 2 shows the variation of junction capacitance C_(j) of a junctiondiode as a function of temperature;

FIG. 3 shows a simplified circuit diagram of a present device accordingto a first embodiment of the present disclosure;

FIG. 4 shows the plot of electrical signals in the circuit of FIG. 3;

FIG. 5 shows a simplified circuit diagram of a present device accordingto a second embodiment of the present disclosure;

FIG. 6 shows the plot of electrical signals in the circuit of FIG. 5;

FIG. 7 shows the plot of the output voltage V_(o) of the circuit of FIG.5; and

FIG. 8 shows a possible implementation of a temperature sensor accordingto the embodiments of FIGS. 3 and 5.

DETAILED DESCRIPTION

Present sensors according to embodiments of the present disclosureexploit the dependence upon temperature of the capacitance of a reversebiased PN-junction diode.

In fact, as is known, the contact potential (or built-in voltage) V_(bi)of a reverse biased PN diode is given by:

$\begin{matrix}{{V_{bi}(T)} = {\frac{k \cdot T}{q} \cdot {\ln( \frac{N_{A} \cdot N_{D}}{n_{i}(T)} )}}} & (1)\end{matrix}$where K is Boltzmann's constant, T is the temperature in degrees Kelvin,q is the charge of the electron, N_(A) is the concentration of acceptoratoms, N_(D) is the concentration of donor atoms, and n_(i)(t) is theconcentration of the intrinsic carriers in the PN diode. In particular,the concentration n_(i) of the intrinsic carriers depends upon thetemperature T on the basis of Eq. (2):

$\begin{matrix}{n_{i}^{2} = {0.961 \cdot 10^{33} \cdot T^{3} \cdot e^{\frac{- {E_{Geff}{(T)}}}{k \cdot T}}}} & (2)\end{matrix}$where E_(Geff) is the energy gap of the material used for integration ofthe diode. In a PN diode, by applying a reverse voltage V_(d) thereto, acharge Q_(j) is stored on the junction:

$\begin{matrix}{{Q_{j}(T)} = \sqrt{2 \cdot q \cdot ɛ_{S} \cdot \frac{N_{D} \cdot N_{A}}{N_{D} + N_{A}} \cdot \lbrack {{V_{bi}(T)} - V_{d}} \rbrack}} & (3)\end{matrix}$where ε_(s) is the dielectric constant of the semiconductor. As may benoted, the accumulated charge depends upon the temperature through thecontact potential V_(bi), as well as upon the reverse voltage V_(d).

The junction capacitance C_(j) of the diode is thus:

$\begin{matrix}{{C_{j}( {T,{Vd}} )} = {{A_{D} \cdot \frac{{dQ}_{j}( {T,V} )}{dV}} = {A_{D} \cdot \frac{{Q_{j}( {T,{Vd}} )} - {Q_{j}( {T,{{Vd} - {dV}}} )}}{dV}}}} & (4)\end{matrix}$where A is the area of the PN junction.

In practice, a PN diode formed in a silicon substrate has a junctioncapacitance depending both upon the biasing voltage and the temperature,as illustrated respectively in FIG. 1 (with a solid line) and in FIG. 2,calculated respectively at constant temperature (T=25° C.) and atconstant reverse biasing voltage (V_(r)=0.625V). FIG. 1 also shows witha dashed line the plot of the junction capacitance determined using moreaccurate calculations.

In particular, as may be noted from FIG. 2, in temperature rangeswherein integrated circuits normally operate, the junction capacitanceC_(j) has an approximately linear plot as a function of temperature.Consequently, to a first approximation, the reading of the junctioncapacitance C_(j) of a reverse biased PN diode has a relation of directproportionality with the local temperature, and reading of the junctioncapacitance and/or of its variation supplies direct information on thetemperature or on the temperature variation.

FIG. 3 shows a temperature sensor 1 that exploits the principleexplained above.

In detail, the temperature sensor 1 comprises a sensor input 2 suppliedwith a sensor excitation signal, i.e., the timed biasing voltage DRH. Asensing diode 3, of a PN-junction type, has its cathode coupled to thesensor input 2 and its anode coupled to an inverting input 7 of anoperational amplifier 4. A reference capacitor 5, having a referencecapacitance C_(R), is coupled between the sensor input 2 and anon-inverting input 8 of the operational amplifier 4. The referencecapacitance C_(R) is chosen so as to have the same value as the junctioncapacitance C_(j) at room temperature.

The inputs 7, 8 of the operational amplifier 4 are both coupled to afirst reference potential line 10, set at a first common mode potentialV_(CMin), through a respective input switch 11. The input switches 11are controlled by a same reset signal R.

The operational amplifier 4 is of a fully differential type, has a pairof outputs 15, 16 and has a capacitive feedback formed by a first and asecond feedback capacitors 17, 18, which have the same feedbackcapacitance C_(i). In detail, the first feedback capacitor 17 is coupledbetween the first output 15 and the inverting input 7, and the secondfeedback capacitor 18 is coupled between the second output 16 and thenon-inverting input 8 of the operational amplifier 4. The outputs 15, 16of the operational amplifier 4 are further coupled to a second referencepotential line 20, set at a second common mode potential V_(CMout),through a respective output switch 21. The output switches 21 arecontrolled by the reset signal R.

A timed biasing voltage DRH is supplied on the input 2 and switchesbetween a low value (for example, 0.625 V) and a high value V_(DRH) (forexample, 1.25V). In particular, the low value is in any case positivefor keeping the sensing diode 3 (which has its anode coupled to thevirtual ground on the inverting input 7 of the operational amplifier 4)reverse biased in all the sensing phases, and the high value is chosenfor generating a voltage step ΔV of a preset value, as explained indetail hereinafter.

In practice, in the temperature sensor 1 of FIG. 3, the sensing diode 3and the reference capacitor 5 form a sensing element 25 and theoperational amplifier 4, with the capacitive feedback, forms a switchedcapacitor differential amplifier stage of a known type and widely used,for example, in reading of MEMS structures.

The operational amplifier 4 and the relevant feedback network 17, 18,11, 21 may be incorporated in an ASIC (Application Specific IntegratedCircuit) 28.

The sensing diode 3 and the reference capacitor 5 may be formed in asemiconductor material chip, as described in greater detail withreference to FIG. 8.

The outputs 15 and 16 of the operational amplifier 4 are coupled to aprocessing stage 30, generally external to the temperature sensor 1 butpossibly also integrated in the ASIC. The processing stage 30 maycomprise circuits for amplification of the output voltage V_(o) and foranalog-to-digital conversion.

Finally, a timing stage 31 generates biasing/timing signals for thetemperature sensor 1 and for the processing stage 30, such as the resetsignal R, the timed biasing voltage DRH, and a reading acquisitionsignal S for the processing stage 30.

Sensing of the output voltage V_(o) of the operational amplifier 4 isobtained according to the timing, illustrated in FIG. 4, which includesthe succession of a reset phase and a sensing phase that follow eachother in an acquisition period T₁ equal to one half of a sensing periodT₂=2·T₁, as described in detail hereinafter. In particular, the resetsignal R and the reading acquisition signal S have the same period, butdifferent duty cycle. For this reason, hereinafter the sensing period T₂is considered as being divided into two half periods T₁₁ and T₁₂,corresponding to two successive periods of the acquisition period T₁.

Reset Phase—First Half Period

At instant t₀ the reset signal R switches to the high state, causing theinput switches 11 and of the output switches 21 to switch off.Consequently, the inputs 7, 8 of the operational amplifier 4 are coupledto the first common mode potential V_(CMin) (for example, 0.625 V, i.e.,to the low value of the timed biasing voltage DRH), and the outputs 15,16 of the operational amplifier 4 are coupled to the second common modepotential V_(CMout) (for example, 1 V), thus resetting the operationalamplifier 4.

In this step, the timed biasing voltage DRH (for example, 0.625 V) islow, as likewise is the reading acquisition signal S. The output voltageV_(o) is thus not acquired by the signal processing stage 30.

Next, at instant t₁, the reset signal R switches to the low state,causing opening of the input and output switches 11, 21 and causing theinput nodes 7, 8 and output nodes 15, 16 of the operational amplifier 4to be independent. The timed biasing voltage DRH is low, as is thereading acquisition signal S.

The step t₁-t₂ may be adopted in case of use of the correlated doublesampling (CDS) technique. During this step, in fact, with the techniquereferred to, offset sampling is carried out, which may then besubtracted during the sensing phase. In this way, it is possible toreduce the offset.

Sensing Phase—First Half Period T₁₁

At the instant t₂, the timed biasing voltage DRH has a rising edge andreaches a value that reverse biases the sensing diode 3, for example, at1.25 V. In this condition, neglecting possible losses, a reverse currentflows in the sensing diode 3. Further, a reference current flows in thereference capacitor 5. There is thus a charge displacement Q₁ (accordingto the law in Eq. (3), where V_(d) is here the timed biasing voltageDRH) from the sensing diode 3 to the operational amplifier 4 and acharge displacement Q₂ (according to the law Q=C/ΔV, where ΔV is thestep of the timed biasing voltage DRH) from the reference capacitor 5 tothe operational amplifier 4. As a result of the half bridgeconfiguration of the sensing element 25 and of the feedback network ofthe amplifier 26, the latter is traversed by a differential charge Q₂-Q₁that is a function of the amplitude of the step ΔV of the timed biasingvoltage DRH, of the capacitance difference ΔC (difference between thejunction capacitance C_(j) of the sensing diode, and the capacitanceC_(R) of the reference capacitor 5), and of the capacitances C_(i) ofthe feedback capacitors 17, 18.

Consequently, between the outputs 15 and 16 of the operational amplifier4 there is an output voltage V_(o)

$\begin{matrix}{{V_{o}(t)} \propto {\frac{\;{C}}{C_{i}}\Delta\; V}} & (5)\end{matrix}$which is acquired by the signal processing stage by virtue of the highvalue of reading acquisition signal S.

In this connection, since the time plot of the output voltage V_(o) hasa transient step, acquisition of the output voltage V_(o) is made duringthe subsequent steady state step regime, and the involved time iscalculated taking into account the bandwidth of the operationalamplifier 4.

This step terminates at instant t₃, where a new period T₁ of the readingacquisition signal S and of the reset signal R and the second halfperiod T₁₂ of the sensing signal DRH start.

Reset Phase—Second Half Period T₁₂

At instant t₃, the reset signal R switches to high, and the readingacquisition signal S switches to low. The operational amplifier 4 isthus reset again, in a way generally similar to what described for thereset phase of the first half period T₁₁, with the only difference that,now, the timed biasing voltage DRH is high. This value does not,however, affect the reset phase since, as before, the operationalamplifier 4 is reset, and the output voltage V_(o) is not acquired bythe signal processing stage 30.

At instant t₄, the reset signal R switches again to low.

Sensing Phase—Second Half Period T₁₂

At the instant t₅, the timed biasing voltage DRH has a falling edge,thus causing a charge displacement opposite to that of the sensing phasein the first half period. Consequently, the output voltage V_(o) has avalue

$\begin{matrix}{{V_{o}(t)} \propto {\frac{C}{C_{i}}\Delta\; V}} & (6)\end{matrix}$with a opposite sign to Eq. (5), since in this half period T₁₂ the stepΔV of the timed biasing voltage DRH is a down step and is equal to−V_(DRH).

Also in this step, the value of the output voltage V_(o) is acquiredfrom the signal processing stage 30 thanks to the high value of thereading acquisition signal S. The signal processing stage 30 thusmodifies the sign of the output voltage V_(o) in one of the two halfperiods T₁₁ and T₁₂, in a way synchronized with the sensing period T₂.

This step terminates at instant t₃, where a new period T₁ of the readingacquisition signal S and of the reset signal R starts, as well as a newperiod T₂ of the sensing signal DRH.

The solution illustrated in FIG. 3 thus enables detection of theabsolute temperature in the area of the sensing diode 3 on the basis ofthe variation of its reverse junction capacitance, using a simplevoltage biased capacitive bridge. This solution presents the advantageof having a zero current consumption and a very simple structure, whichdoes not require use of MEMS capacitive structures.

However, the sensing diode 3 intrinsically presents a current leakagethat, in some situations, may reduce reading precision in an undesirableway.

In fact, the aforesaid current leakage of the sensing diode 3 determinesa variation of the charge Q₁. In fact, during the sensing phase, thesensing diode 3 undergoes a displacement of charge equal to Q₁−Q_(L),where Q_(L) is the charge due to the leakage current. A more preciseapproximation of the output voltage V_(o) is thus the following:

$\begin{matrix}{{V_{o}(t)} \propto {{\frac{C}{C_{i}}\Delta\; V} + {I_{L} \cdot C_{i} \cdot \frac{T_{1}}{2}}}} & (7)\end{matrix}$where I_(L) is the leakage current of the sensing diode 3.

FIG. 5 shows an embodiment wherein the leakage current I_(L) of thesensing diode 3 is compensated.

In particular, the temperature sensor of FIG. 5 has the same basicstructure as the temperature sensor of FIG. 3, where a compensationdiode 40 and a symmetry capacitor 41, having a capacitance equal to thecapacitance C_(R) of the reference capacitor 5, have been added. Theelements in common with those of FIG. 3 are thus designated by the samereference numbers.

In detail, the compensation diode 40 has its anode coupled to thenon-inverting input 8 of the operational amplifier 4 and its cathodecoupled to a third reference potential line 43, set at a non-constantbiasing potential V_(STB), and the symmetry capacitor 41 is coupledbetween the inverting input 7 of the operational amplifier 4 and thethird reference potential line 43.

The biasing potential V_(STB) switches between two positive values, forexample between 0 V and 2.5 V, to be surely higher than the potential onthe inputs 7 and 8 of the operational amplifier 4 and keep thecompensation diode 40 reverse biased.

Two current generators 45 are also illustrated in FIG. 5 and representthe leakage currents I_(L) of the sensing diode 3 and of thecompensation diode 40. In practice, the sensing diode 3, the referencecapacitor 5, the compensation diode 40, and the symmetry capacitor 41form a capacitive bridge 44.

As illustrated in the timing diagram of FIG. 6, where the signals R,DRH, S have the same meaning as in FIG. 4, and the switching instantst₀-t₅ correspond to the above, the biasing potential V_(STB) switches atthe rising edge of the reset signal R, thus at frequency f₂=1/T₂ equalto one half of the frequency (f₁=1/T₁) of the reset signal R and equalto the frequency of the sensing signal DRH. It follows that the effectsof switching of the biasing potential V_(STB) do not affect theoperational amplifier 4 since, during the reset phase, the inputs 7, 8of the latter are coupled to the first common mode potential V_(CMin).Furthermore, the biasing potential V_(STB) has the same sign as thetimed biasing voltage DRH during the sensing phase; namely, it ispositive with respect to the virtual ground on the inputs 7 and 8 of theoperational amplifier 4 for keeping the compensation diode 40 reversebiased in all the sensing phases and to have a voltage step ΔV equal tothe step of the sensing signal DHR in order to keep the compensationdiode 40 in the same operating conditions as the sensing diode 3.

In this way, during the sensing phase (both in the first half period T₁₁and in the second half period T₁₂ of the sensing signal DRH), thefeedback capacitors 17 and 18 are traversed by the following currents:

a current due to the differential charge generated by the switching edgeof the timed biasing voltage DRH and depending on the difference ofcapacitance in the capacitive bridge 44;

a leakage current I_(L1) in the sensing diode 3; and

a leakage current I_(L2) in the compensation diode 40.

Consequently, the output voltage V_(o) of the temperature sensor of FIG.5 may be expressed as follows:

$\begin{matrix}{{V_{o}(t)} \propto {{\frac{C}{C_{i}}\Delta\; V} + {I_{L\; 1} \cdot C_{i} \cdot \frac{T_{1}}{2}} - {I_{L\; 2} \cdot C_{i} \cdot \frac{T_{1}}{2}}}} & (8)\end{matrix}$

By manufacturing the compensation diode 40 in the same way and with thesame parameters as the sensing diode 3, due also to the same reversebiasing of the diodes 3 and 40, they generate leakage currents I_(L1)and I_(L2) that are the same so that in Eq. (8) the two contributions ofthe leakage currents I_(L1) and I_(L2) cancel out, and the outputvoltage V_(o) may be expressed again by Eq. (6).

The effect of cancelling out of the leakage currents in the sensingdiode 3 is visible in the simulation of FIG. 7, which reproduces thevoltages on the non-inverting output 15 and on the inverting output 16of the operational amplifier 4 in case of compensation of the leakagecurrent of the sensing diode 3 (solid lines) and in the case withoutcompensation (dashed lines).

As may be noted, the output voltage V_(o) of the operational amplifier 4without compensation has a reading error proportional both to theleakage current and to the integration time. Instead, the output voltageV_(o) with compensation, after a transient, is insensitive to the aboveparameters.

FIG. 8 shows a possible implementation of the compensation diode 40 andof the symmetry capacitor 41. In the example illustrated, thetemperature sensor 1 is formed in a chip 60 of semiconductor material,such as silicon, having a substrate 50 of a P type, accommodates a well51 of an N type, which forms the cathode K of the compensation diode 40.The well 51 in turn accommodates a tap 52, of a P type, forming theanode A of the compensation diode 40.

An insulating layer 55 extends over the substrate 50 and accommodatestwo metal regions 56, 57, arranged on top of each other and formed, forexample, in two different metallization levels of the chip 60. The metalregions 56, 57 form, together with the portion of the insulating layer55 arranged in between, the reference capacitor 5.

The compensation diode 40 and the symmetry capacitor 41 may be formed ina similar way.

The described temperature sensor comprises only a few simple componentsof a capacitive type (sensing diode 3, reference capacitor 5, possibly acapacitive bridge 44) that may easily be integrated and require only asmall area, which cooperate with a sensing network (operationalamplifier 4 and corresponding feedback network) that may be manufacturedusing standard CMOS technology. The sensor has a zero d.c. biasingvoltage, and thus a low current consumption.

The temperature sensor 1 may be compensated with respect to the currentleakages by a few simple components (compensation diode 40, symmetrycapacitor 41), thus supplying a particularly precise output.

Finally, it is clear that modifications and variations may be made tothe embodiments of a temperature sensor described and illustratedherein, without thereby departing from the scope of the presentdisclosure. In particular, the switched capacitor differential amplifier26 may be replaced by another type of sensing circuit, and/or be formedin a different way from what illustrated, for example be formed as anon-fully differential amplifier.

The various embodiments described above can be combined to providefurther embodiments. All of the U.S. patents, U.S. patent applicationpublications, U.S. patent applications, foreign patents, foreign patentapplications and non-patent publications referred to in thisspecification and/or listed in the Application Data Sheet areincorporated herein by reference, in their entirety. Aspects of theembodiments can be modified, if necessary to employ concepts of thevarious patents, applications and publications to provide yet furtherembodiments.

These and other changes can be made to the embodiments in light of theabove-detailed description. In general, in the following claims, theterms used should not be construed to limit the claims to the specificembodiments disclosed in the specification and the claims, but should beconstrued to include all possible embodiments along with the full scopeof equivalents to which such claims are entitled. Accordingly, theclaims are not limited by the disclosure.

The invention claimed is:
 1. A sensing method, comprising: applying areverse biasing voltage across a junction diode, the junction diodehaving an anode and a cathode, and the junction diode having a junctioncapacitance that varies as a function of temperature; applying a voltageto the cathode that is positive with respect to a voltage on the anode;switching the voltage applied to the cathode between a low value and ahigh value to apply a reverse voltage step across the anode and cathodeof the junction diode; detecting a change in the junction capacitance ofthe junction diode as a function of temperature in response to thereverse voltage step; and generating an output signal having a valuebased on the detected change in junction capacitance, the output signalindicating a temperature of the junction diode.
 2. The method of claim1, wherein detecting a change in the junction capacitance of thejunction diode comprises detecting a charge flowing through the junctiondiode responsive to a change in the reverse biasing voltage.
 3. Themethod of claim 2, wherein detecting a change in the junctioncapacitance of the junction diode further comprises detecting a chargeflowing through a reference capacitor responsive to the change in thereverse biasing voltage, the reference capacitor having a value that isapproximately equal to the value of the junction capacitance of thejunction diode at room temperature.
 4. The method of claim 3, whereindetecting the charge flowing through the junction diode and thereference capacitor comprises: applying a first reverse bias voltageacross the junction diode and the reference capacitor; generating afirst value of the output signal responsive to applying the firstreverse bias voltage across the junction diode and reference capacitor;applying a second reverse bias voltage across the junction diode and thereference capacitor, the second reverse bias voltage being differentthan the first reverse bias voltage; generating a second value of theoutput signal responsive to applying the second reverse bias voltageacross the junction diode and reference capacitor; and detecting thetemperature of the junction diode based on the first and second valuesof the output signal.
 5. The method of claim 4, further comprisingcompensating for a leakage current through the junction diode.
 6. Themethod of claim 5, wherein the anode of the junction diode is coupled toa first node, and wherein compensating for the leakage current throughthe junction diode further comprises providing a symmetry capacitance onthe first node.
 7. The method of claim 6, wherein the referencecapacitor is coupled to a second node, and wherein compensating for theleakage current through the junction diode further comprises coupling ananode of a compensation diode to the second node.
 8. A method,comprising: providing a voltage to reverse bias a PN junction of ajunction diode, the PN junction having a junction capacitance; providinga reverse bias voltage change across the PN junction; detecting a valueof the junction capacitance in response to the reverse bias voltagechange, the value of the junction capacitance being a function of atemperature of the PN junction; and generating an output signal based onthe detected junction capacitance, the output signal indicating atemperature of an environment containing the junction diode.
 9. Themethod of claim 8, wherein detecting a value of the junction capacitanceof the PN junction comprises detecting a charge flowing through the PNjunction responsive to a change in the voltage provided to reverse biasthe PN junction.
 10. The method of claim 8, further comprisingcompensating for a leakage current through the PN junction.
 11. Themethod of claim 10, wherein the leakage current through the PN junctionis compensated with a compensation current corresponding to a leakagecurrent of a compensation diode, the compensation current beingapproximately equal to the leakage current through the PN junction. 12.The method of claim 8, further comprising charging first and secondnodes to a first reference voltage, the first node being coupled to thejunction diode, the second node being coupled to a reference capacitor.13. The method of claim 12, further comprising, after charging the firstnode to the first reference voltage, providing a charge flowing throughthe PN junction to the first node responsive to a change in the voltageprovided to reverse bias the PN junction.
 14. The method of claim 13,further comprising, after charging the second node to the firstreference voltage, providing a charge flowing through the referencecapacitor to the second node.
 15. The method of claim 14, furthercomprising providing a compensation current to the second node tocompensate for a leakage current through the PN junction.
 16. The methodof claim 15 further comprising providing an additional capacitance onthe first node having value that is approximately equal to a capacitanceof the reference capacitor.
 17. A sensing method, comprising: applying areverse biasing voltage across a junction diode, the junction diodehaving an anode coupled to a first node and having a cathode coupled toa second node, and the junction diode having a junction capacitance thatvaries as a function of temperature; charging the first node to a firstreference voltage; applying a biasing voltage to the second node that ispositive with respect to the first biasing voltage on the first node;switching the biasing voltage between a low value and a high value toapply a reverse voltage step across the anode and cathode of thejunction diode; providing a charge from the second node through thejunction diode to charge the first node in response to the biasingvoltage switching between the low and high values; charging a third nodeto the first reference voltage; providing a charge from the second nodethrough a reference capacitance to the third node in response to thebiasing voltage switching between the low and high values; andgenerating an output signal indicating temperature of the junction diodebased on voltages on the first and second nodes.
 18. The method of claim17, wherein generating the output signal comprises taking a differencebetween a voltage on the first node and a voltage on the third node. 19.The method of claim 17, wherein the charging of the first and thirdnodes occurs during a reset phase of operation, and the providing of thecharge from the second node through the junction diode to charge thefirst node and the providing of the charge from the second node throughthe reference capacitance to the third node, and the generating of theoutput signal, occur during a sensing phase of operation.
 20. The methodof claim 17, further comprising providing a compensating capacitance onthe first node, the compensating capacitance having a value that isapproximately equal to a value of the reference capacitance.
 21. Themethod of claim 20, further comprising providing a leakage current tothe third node, the leakage current being approximately equal to aleakage current through the junction diode.